In general, voltage-mode control switching regulators have been widely used. The voltage-mode control switching regulators perform PWM control on a switching element in accordance with a voltage difference between an output voltage and a reference voltage, thereby stabilizing the output voltage. However, since the voltage-mode control switching regulators detect a returned signal from the output voltage, their response speed to fluctuation in the output voltage is slow and the phase compensation of an error amplification circuit which amplifies the voltage difference between the output voltage and the reference voltage becomes complicated.
As a technology to solve these deficiencies, current-mode control switching regulators have been used in many cases. However, it is known that, when the on-duty cycle of PWM control exceeds 50%, the current mode control switching regulators cause subharmonic oscillation and go out of control. Therefore, slope compensation is performed on PWM control to prevent subharmonic oscillation.
FIG. 1 is a circuit diagram showing an example of a current-mode control switching regulator having a circuit that performs such slope compensation and shows a step-down switching regulator.
In FIG. 1, when a switching transistor 105 is turned on, power is supplied to an inductor 104, a smoothing capacitor 102, and a load 101. When the switching transistor 105 is turned off, the energies accumulated in the inductor 104 and the smoothing capacitor 102 are supplied to the load 101. A current/voltage conversion circuit 106 has impedance Rsense and outputs a converted voltage Vsense (=Rsense×iL) obtained by converting the voltage of a current iL fed to the inductor 104 with the impedance Rsense.
Furthermore, an oscillation circuit 110 generates and outputs a predetermined reference clock signal CLK and a predetermined sawtooth wave voltage Vramp. An accumulator 108 performs slope compensation by adding the sawtooth wave voltage Vramp to the converted voltage Vsense and outputs the result to the non-inverting input terminal of a PWM comparator 107 as a slope voltage Vs. An error amplification circuit 115 amplifies a voltage difference between a reference voltage Vref and a divided voltage Vfb obtained by dividing an output voltage Vout and outputs a generated error voltage Ve to the inverting input terminal of the PWM comparator 107. The PWM comparator 107 compares the error voltage Ve with the slope voltage Vs. When the error voltage Ve exceeds the slope voltage Vs, the PWM comparator 107 resets an RS latch circuit 112 to turn off the switching transistor 105. Therefore, the peak current value of the inductor current iL depends on the error voltage Ve.
In regulating the output voltage Vout, the PWM comparator 107 reduces the error voltage Ve to reduce the output voltage Vout when the divided voltage Vfb is larger than the reference voltage Vref. Furthermore, the PWM comparator 107 increases the error voltage Ve to increase the output voltage Vout when the divided voltage Vfb is smaller than the reference voltage Vref.
In order to prevent the subharmonic oscillation described above, it is necessary to perform slope compensation so that the inclination of the slope voltage Vs becomes one-half or more of that of the inductor current iL when the switching transistor 105 is turned off.
Specifically, in FIG. 1, assuming that the inductance of the inductor 104 is L, the inclination diL/dt of the inductor current iL when the switching transistor 105 is turned on is obtained by the following formula (a). Furthermore, the inclination diL/dt of the inductor current iL when the switching transistor 105 is turned off is obtained by the following formula (b).diL/dt=(Vin−Vout)/L  (a)diL/dt=−Vout/L  (b)
When the inclination of the sawtooth wave voltage Vramp is slope compensation Iramp, the slope compensation at this time is obtained by the following formula (c).Iramp>Vout/2/L×Rsense  (c)
Note that in a step-up switching regulator, the formulae (a), (b), and (c) are represented by the following formulae (d), (e), and (f), respectively.diL/dt=Vin/L  (d)diL/dt=−(Vout−Vin)/L  (e)Iramp>(Vout−Vin)/L/2×Rsense  (f)
As described above, the slope compensation Iramp can be represented using the variables of the output voltage Vout and the input voltage Vin. When the input voltage Vin and the output voltage Vout are constant values, no problem occurs. However, the input voltage Vin and the output voltage Vout generally fluctuate in a wide range. Therefore, when the slope compensation Iramp is set to a fixed value, the slope compensation Iramp is required to be set to the maximum value within the fluctuation range of the expected input voltage Vin and the output voltage Vout. However, although subharmonic oscillation can be prevented when excessive slope compensation is performed, the advantages of current feedback are lost. As a result, the current-mode control switching regulator operates like the voltage-mode switching regulator to degrade its controllability. In order to deal with this, a slope control amount is determined in accordance with an input/output voltage to properly perform slope compensation in a wide input/output voltage range (see, e.g., Patent Document 1).
Patent Document 1: JP-2006-33958
However, in this case, since the slope compensation amount is changed in accordance with an input voltage and an output voltage, the circuit becomes complicated. Furthermore, since the general-purpose IC of the switching regulator generally has an external resistor for generating the divided voltage obtained by dividing an output voltage and cannot monitor the output voltage, it cannot perform slope compensation in accordance with the output voltage.
On the other hand, the inductance L selected by a designer is changed as the operating frequency of the switching regulator changes. Therefore, when the slope compensation Iramp is set to a fixed value, it is required to be set to the maximum value within the fluctuation range of the expected input voltage Vin, the output voltage Vout, and the inductance L. For example, when the oscillation frequencies of the oscillation circuit are 2 MHz, 1 MHz, 500 kHz, and 300 kHz, the selected inductances L are 2.2 μH, 4.7 μH, 10 μH, and 15 μH, respectively, which are inversely proportional to the oscillation frequencies. However, as described above, although subharmonic oscillation can be prevented when excessive slope compensation is performed, the advantages of current feedback are lost. As a result, the current-mode control switching regulator operates like the voltage-mode switching regulator to degrade its controllability. However, the known switching regulator cannot perform the slope compensation in accordance with the inductance L.